The present invention relates to a switching power circuit equipped with a power factor improving circuit.
Previously to date the present applicant proposed a variety of power circuits each having a resonance type converter on its primary side, and also other various power circuits each having a power factor improving circuit to achieve improvement of the power factor for a resonance type converter.
FIG. 6 is a circuit diagram showing an exemplary switching power circuit of a configuration based on the invention filed previously by the present applicant. This power circuit is equipped with a power factor improving circuit to attain improvement of the power factor for a self-excited voltage resonance type switching converter.
In the switching power circuit shown in this diagram, there are provided a common mode choke coil CMC and an across capacitor CL which constitute a noise filter to remove the common mode noise with regard to an alternating current power AC.
The alternating current power AC is full-wave rectified by a bridge rectifier circuit Di consisting of four diodes, and the rectified output is supplied to charge a smoothing capacitor Ci via a power factor improving circuit 20. The circuit configuration of the power factor improving circuit 20 and the operation thereof will be described later.
In this diagram, the voltage resonance type switching converter has a switching element Q1 consisting, for example, of a high-voltage withstanding bipolar transistor. That is, this switching converter is in a single end form.
The base of the switching element Q1 is connected to the positive side of the smoothing capacitor Ci via a starting resistor RS, so that a base current at the start is obtained from a rectifier smoothing line. The base of the switching element Q1 is connected also to a switching drive circuit 2.
The switching drive circuit 2 consists of a self-excited oscillation driver for driving the switching element Q1 by self-excitation, and a switching frequency controller for stabilizing the voltage by varying the oscillation frequency (i.e., switching frequency) in the self-excited oscillation driver.
In a specific configuration of such switching drive circuit 2, as seen in the various power circuits proposed previously by the present applicant, the self-excited oscillation driver consists, for example, of a resonance circuit composed of a driving coil and a resonance capacitor, and a detection coil for transferring the alternating voltage to the driving coil. Although not shown here, the detection coil is connected practically in series to a primary winding N1 for example. That is, the switching element Q1 is driven through switching by the resonance output of the resonance circuit in the self-excited oscillation driver, and the resonance frequency thereof is used as a switching frequency.
The switching frequency controller has a structure adapted to vary the resonance frequency. For this purpose, a control transformer PRT is provided to vary the inductance of the driving coil for example. In this control transformer PRT, the driving coil and the detection coil for example are transformer-coupled to each other, and a control coil is wound in such a manner that the winding direction thereof is not coincident with that of the driving coil and the detection coil. A DC control current outputted from the control circuit 1 is supplied to the control coil.
In the control circuit 1, a control current of a level corresponding to a secondary side DC output voltage Eo is supplied to the control coil. In the control transformer PRT, the inductance of the driving coil is varied in accordance with the level of the control current flowing in the control coil. As the inductance of the driving coil is thus varied, the resonance frequency of the self-excited oscillation driver, i.e., the switching frequency, is also varied under control.
A detailed description will be given later on a constant voltage regulating action executed with such switching frequency control.
The collector of the switching element Q1 is connected to the positive terminal of the smoothing capacitor Ci via the primary winding N1 of the insulating converter transformer PIT, and the emitter thereof is grounded.
In this case, a clamp diode DD is connected between the collector and emitter of the switching element Q1, thereby forming a path where a damper current flows at the off-time of the switching element Q1.
A first resonance capacitor Cr constitutes a parallel resonance circuit in combination with a second resonance capacitor Cr1 in an undermentioned power factor improving circuit 2 and principally with the leakage inductance of the primary winding N1 of the insulating converter transformer PIT. Due to the action of this parallel resonance circuit, the switching operation of the switching element Q1 is performed in a voltage resonance mode. And the end voltage VCP between the collector and emitter of the switching element Q1 is obtained in a waveform of sinusoidal pulses during the off-period of the switching element.
The insulating converter transformer PIT transfers the switching output of the switching element Q1 to the secondary side.
As shown in FIG. 12, the insulating converter transformer PIT has an EE-shaped core where E-shaped cores CR1 and CR2 composed of ferrite for example are combined with each other in such a manner that magnetic legs thereof are opposed mutually, and the primary windings N1 and the secondary windings N2 thereof are coiled in a split state respectively by the use of a split bobbin B with regard to the center magnetic leg of the EE-shaped core. And a gap G is formed to the center magnetic leg as shown in the diagram, whereby loose coupling is attained with a required coupling coefficient.
The gap G can be formed by shaping the center magnetic leg of each of the E-shaped cores CR1 and CR2 to be shorter than the two outer magnetic legs thereof. The coupling coefficient k is set as, e.g., k≈0.85 suited to attain loose coupling, hence avoiding a saturated state correspondingly thereto.
One end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1, while the other end thereof is connected to the positive side (rectified smoothed voltage Ei) of the smoothing capacitor Ci.
On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, since a secondary side parallel resonance capacitor C2 is connected to the secondary winding N2, a parallel resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonance capacitor C2. The alternating voltage thus produced in the secondary winding N2 is turned into a resonance voltage in the parallel resonance circuit. That is, a voltage resonance operation is performed on the secondary side.
More specifically, this power circuit has, on its primary side, a parallel resonance circuit to execute the switching operation in a voltage resonance mode, and also has, on its secondary side, another parallel resonance circuit to perform a voltage resonance operation in the rectifier circuit. In this specification, the switching converter of a configuration equipped with resonance circuits on its primary and secondary sides as mentioned above will be referred to as xe2x80x9ccomposite resonance type switching converterxe2x80x9d.
In the secondary side parallel resonance circuit formed as described, a center tap is provided for the secondary winding N2, and there is also provided a half-wave rectifier circuit consisting of a rectifier diode D0 and a smoothing capacitor Co. This half-wave rectifier circuit receives an input resonance voltage supplied from the secondary side parallel resonance circuit and delivers a DC output voltage Eo therefrom.
In the insulating converter transformer PIT, the mutual inductance M regarding the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 becomes either +M or xe2x88x92M depending on the relation of the polarities (winding directions) of the primary winding N1 and the secondary winding N2 to the connection of the rectifier diode D0.
For example, if the connection is in a state of FIG. 13A, the mutual inductance becomes +M (forward mode). Meanwhile, if the connection is in a state of FIG. 13B, the mutual inductance becomes xe2x88x92M (flyback mode).
Applying the above to the secondary side operation of the power circuit shown in FIG. 6, when the alternating voltage obtained in the secondary winding N2 is positive for example, it is supposed that the operation with the rectified current flowing in the rectifier diode D0 is performed in the +M (forward) mode. That is, in the power circuit of FIG. 6, the mutual inductance functions in the +M (forward) mode every time the alternating voltage obtained in the secondary winding is turned to positive/negative.
In this configuration, the power is supplied to the load increased by the action of the secondary side parallel resonance circuit, so that the power supplied to the load is also increased correspondingly thereto to consequently enhance the increase rate of the maximum load power.
Such correspondence to the load condition can be realized due to the improved situation where a saturated state is not reached readily because of the loose coupling attained by a required coupling coefficient with the gap G formed in the insulating converter transformer PIT, as explained previously with reference to FIG. 12. For example, in case the gap G is not existent in the insulating converter transformer PIT, the operation will be abnormal with a high probability as the insulating converter transformer PIT is placed in its saturated state during the flyback, whereby proper execution of the rectification is rendered considerably difficult.
In the circuit of FIG. 6 where the switching frequency is varied for constant voltage control, the switching frequency variation is executed under control to vary the on-time of the switching element Q1 while maintaining the off-time thereof fixed. That is, in this power circuit, the constant voltage control is performed to vary the switching frequency under control to thereby control the resonance impedance with regard to the switching output, and simultaneously the conduction angle control (PWM control) of the switching element in the switching period is also performed. Such composite control operation is realized in a single set of control circuitry.
In executing such switching frequency control, when the secondary side output voltage is increased due to a trend of reduction of the load for example, the secondary side output is suppressed under control by raising the switching frequency.
The power factor is improved by the power factor improving circuit 20.
As shown in this diagram, the power factor improving circuit 20 has a series connected circuit of a choke coil Ls and a fast recovery diode D2 inserted between the positive output of a bridge rectifier circuit Di and the positive terminal of a smoothing capacitor Ci. The anode of the fast recovery diode D2 is connected to the choke coil Ls, while the cathode thereof is connected to the positive terminal of the smoothing capacitor Ci. The choke coil Ls functions as a load of the switching output fed back as will be described later.
And a filter capacitor CN is connected in parallel to the series connection of the choke coil Ls and the fact recovery diode D2. In this circuit, the choke coil Ls and the filter capacitor CN constitute a normal-mode LC low pass filter which prevents inflow of any switching high-frequency noise into the AC line.
The junction (voltage division point) in the aforementioned series connection of the first parallel resonance capacitor Cr and the second parallel resonance capacitor Cr1 is connected to the junction of the choke coil Ls and the fast recovery diode D2.
It is supposed here that the respective values are selectively set as the first parallel resonance capacitor Cr=8200 pF, second parallel resonance capacitor Cr1=0.027 xcexcF, choke coil Ls=75 xcexcH, and filter capacitor CN=1 xcexcF.
In the power factor improving circuit 20 of such a connected circuit configuration, a resonance pulse voltage Vcp obtained at the off-time during the switching operation of the switching element Q1 is divided by the series connection of the first parallel resonance capacitor Cr and the second parallel resonance capacitor Cr1, and then the divided voltage is impressed in such a manner as to be fed back to the junction of the choke coil Ls and the fast recovery diode D2.
When the resonance pulse voltage Vcp is 600 Vp for example, a voltage of approximately 150 vp obtained through voltage division at 3:1 or so is fed back to the junction of the choke coil Ls and the fast recovery diode D2.
At the timing when the alternating input voltage VAC reaches the vicinity of its positive or negative peak, the fast recovery diode D2 is turned on. At this time, a pulse current having a steep rise waveform comes to flow from the output terminal of the bridge rectifier circuit Di into the smoothing capacitor Ci to charge the same via the choke coil Ls and the fast recovery diode D2.
Meanwhile, in any other period than when the alternating input voltage VAC reaches the vicinity of its positive or negative peak, the fast recovery diode D2 repeats its switching operation in accordance with the resonance pulse voltage fed back as voltage V2, as mentioned. And at the timing when the fast recovery diode D2 is turned off during the switching operation, a parallel resonance current flows in a circuit consisting of the second parallel resonance capacitor Cr1, the choke coil Ls and the filter capacitor CN. Meanwhile, at the timing when the fast recovery diode D2 is turned on, a high-frequency charge current flows in the smoothing capacitor Ci from the alternating input voltage VAC via the choke coil Ls.
In this manner, utilizing the primary side voltage resonance pulse fed back to the rectified current path, the current to be caused to flow in the rectified current path is converted into a high-frequency current to attain an alternating action, hence widening the conduction angle of the alternating input current IAC to consequently improve the power factor.
FIG. 7 graphically shows characteristics of the power circuit having the configuration of FIG. 6, including changes of the power factor derived from load variations and also changes of the DC input voltage (rectified smoothed voltage Ei). This diagram represents a comparison between the characteristics (solid line) of the circuit equipped with the power factor improving circuit 20 of FIG. 6 and the characteristics of another circuit which is not equipped with the power factor improving circuit 20 of FIG. 6.
According to this graphic diagram, it is obvious that, in a range of the load power Po from 0 W to 200 W, the power factor PF is improved more in the circuit configuration with the power factor improving circuit 20 of FIG. 6 than in another circuit configuration without the power factor improving circuit. Particularly in the circuit shown in FIG. 6, there is attained such characteristic that the power factor indicates its peak value when the load power Po is 50 W or so.
It is also seen that the level of the rectified smoothed voltage Ei tends to become higher in accordance with reduction of the load power Po.
FIG. 8 graphically shows change characteristics of the power factor relative to variations of the alternating input voltage VAC and the DC input voltage (rectified smoothed voltage Ei). This diagram also represents a comparison between the characteristics (solid line) of the circuit equipped with the power factor improving circuit 20 of FIG. 6 and the characteristics of another circuit which is not equipped with the power factor improving circuit 20 of FIG. 6.
As shown in this diagram, the characteristic obtained is such that the power factor is lowered in the circuit configuration where none of power factor improvement is performed in accordance with a rise of the alternating input voltage VAC in a range of 80 V to 140 V, while the power factor PF can be enhanced in the circuit of FIG. 6, where the power factor PF is increased with a rise of the alternating input voltage VAC.
There is further achieved such characteristic that the rectified smoothed voltage Ei is raised with a rise of the alternating input voltage VAC.
FIG. 9 shows another example of a switching power circuit constituted on the basis of the invention proposed previously by the present applicant. In this power circuit also, a power factor improving circuit is included to achieve power factor improvement for a self-excited voltage resonance type switching converter. In this diagram, any component parts equivalent to those in FIG. 6 are denoted by the same reference numerals, and a repeated explanation thereof is omitted.
The power circuit shown in this diagram is equipped with a power factor improving circuit 21. In comparison with the aforementioned power factor improving circuit 20 of FIG. 6, this power factor improving circuit 21 is different in the point that the fast recovery diode D2 and the choke coil Ls therein are connected reversely. That is, the anode of the fast recovery diode D2 is connected to the positive output terminal of the bridge rectifier circuit Di, and the cathode thereof is connected to one end of the choke coil Ls. Meanwhile the other end of the choke coil Ls is connected to the positive terminal of the smoothing capacitor Ci. And resonance pulses voltage Vcp, which are obtained through voltage division by a first parallel resonance capacitor Cr and a second parallel resonance capacitor Cr1, are impressed to the junction of the fast recovery diode D2 and the choke coil Ls.
In such circuit configuration also, the fast recovery diode D2 is turned on at the timing when the alternating input voltage VAC reaches the vicinity of its positive or negative peak, and a pulse current having a steep rise waveform comes to flow from the output terminal of the bridge rectifier circuit Di into the smoothing capacitor Ci to charge the same via the fast recovery diode 2 and the choke coil Ls.
In this case, the fast recovery diode D2 is temporarily turned off when the absolute value of the alternating input voltage VAC has lowered to a certain level, and voltage resonance is induced at this time by a parallel resonance circuit consisting of the second parallel resonance capacitor Cr1 and the choke coil Ls. Due to such voltage resonance, a sine-wave pulse voltage is superposed on the cathode potential V2 (divided resonance pulse voltage) of the fast recovery diode D2. Subsequently the fast recovery diode D2 repeats its switching operation in response to the potential difference between the cathode potential V2 and the anode potential V1 of the fast recovery diode D2. And a charge current is caused to flow from the filter capacitor CN into the smoothing capacitor Ci during the on-time of the fast recovery diode D2 in such switching operation. This performance extends the conduction angle of the alternating input current IAC to thereby improve the power factor.
FIG. 10 graphically shows characteristics of the power circuit having the configuration of FIG. 9, including changes of the power factor derived from load variations and also changes of the DC input voltage (rectified smoothed voltage Ei). And FIG. 11 graphically shows change characteristics of the power factor relative to variations of the alternating input voltage and the DC input voltage (rectified smoothed voltage Ei). In view of a description to be given later, these diagrams represent the characteristics in one case where the constant of the second parallel resonance capacitor Cr1 is set to 0.033 xcexcF and in another case where the constant is set to 0.043 xcexcF.
First, as seen from FIG. 10, the power factor PF can be kept over 0.70 when the load power Po is substantially within a practical range of 50 W to 200 W. Regarding the rectified smoothed voltage Ei, the result obtained indicates that the voltage Ei tends to rise in accordance with reduction of the load power Po.
And according to the characteristics shown in FIG. 11, it is seen that the power factor PF can be kept over 0.7 with respect to variations of the alternating input voltage VAC ranging from 80 V to 140 V, and the rectified smoothed voltage Ei rises in accordance with rise of the alternating input voltage VAC.
Although the power factor PF can thus be enhanced by the provision of such power factor improving circuit 20 or 21 as shown in FIG. 6 or 9, it is known that the ripple component superposed on the DC input voltage (rectified smoothed voltage Ei) is increased due to the circuit configuration of the power factor improving circuit 20 or 21 where the switching output is fed back to the rectified current path.
In the circuit of FIG. 6 for example, the ripple component xcex94Ei of 9.2 V, which is superposed on the rectified smoothed voltage Ei in the configuration without the power factor improving circuit 20, increases to 35.3 V in the configuration equipped with the power factor improving circuit 20. Particularly in a no-load state, the ripple component xcex94Ei rises to 26 V or so.
The same result is attainable also in the power circuit of the configuration shown in FIG. 9.
Supposing now that, in the configuration of FIG. 9 for example, the respective values are selectively set as the first parallel resonance capacitor Cr=8200 pF, the second parallel resonance capacitor Cr1=0.027 xcexcF and the choke coil Ls=75 xcexcH, then the power factor PF can be kept over 0.73 with the load power Po ranging from 25 W to 200 W, but the ripple component xcex94Ei increases to 31.8 V.
In the circuit of FIG. 9, if the value of the second parallel resonance capacitor Cr1 is set to 0.033 xcexcF or 0.043 xcexcF for changing the voltage division ratio of the first parallel resonance capacitor Cr and the second parallel resonance capacitor Cr1 to adjust (reduce) the feedback quantity of the voltage resonance pulses, then ripple component xcex94Ei decreases to 25.3 V in the case of Cr1=0.033 xcexcF, or further decreases to 9.1 V in the case of Cr1=0.043 xcexcF.
Thus, it is possible to suppress the ripple component xcex94Ei by reducing the feedback quantity of the voltage resonance pulses. However, if the feedback quantity of the voltage resonance pulses is reduced, the power factor PF is lowered. For example, such characteristic is shown in FIGS. 10 and 11 as well, wherein a higher power factor is obtained in the case of Cr1=0.043 xcexcF than in the case of Cr1=0.033 xcexcF. Therefore, in the circuit of FIG. 9 for example, adjustment is so executed as to suppress the ripple component xcex94Ei and to obtain a practically satisfactory power factor PF with a limit of Cr1=0.043 xcexcF or so. The same may be said with regard to the circuit of FIG. 6 also.
In the circuits of FIGS. 6 and 9, the level of the DC input voltage (rectified smoothed voltage Ei) rises with reduction of the load power Po as shown in FIGS. 7 and 10 respectively, and its rise rate becomes high since the ripple component xcex94Ei increases particularly in accordance with approach to the no-load condition. This signifies an increase of the voltage variation rate in relation to any load variation.
Therefore, in an AC 100 V system, the required withstand voltage of the smoothing capacitor Ci for generating a DC input voltage is 200 V in a case without power factor improvement, but it needs to be 250 V in a configuration to execute power factor improvement. Further in an AC 200 V system, the required withstand voltage is 400 V in a case without power factor improvement, but it needs to be 500 V in a configuration to execute power factor improvement.
As a result, there exists a problem that the smoothing capacitor Ci is rendered dimensionally larger to eventually fail in downsizing the circuit and curtailing the production cost.
The smoothing capacitor Ci consists of an electrolytic capacitor for example, and if the selected withstand voltage of the electrolytic capacitor employed here has a higher value while the capacitance thereof is left unchanged, then its equivalent internal resistance increases to eventually increase also the amount of the self-generated heat. Therefore, the deterioration degree of the electrolytic capacitor due to its secular change becomes greater to consequently lower the reliability correspondingly thereto.
The present invention has been accomplished in view of the problems mentioned above.
According to the one aspect of the present invention, there is provided a switching power circuit comprises rectifying and smoothing means for generating a rectified smoothed voltage out of an input commercial AC power and outputting the same as a DC input voltage; an insulating converter transformer having a gap to obtain a required coupling coefficient for loose coupling and serving to transfer a primary output to a secondary side; switching means for outputting, to the primary winding of the insulating converter transformer, the switching output obtained through on/off control of the DC input voltage by a switching element; and a primary side parallel resonance circuit for actuating the switching means in a voltage resonance mode, wherein the resonance circuit consists at least of a leakage inductance component including the primary winding of the insulating converter transformer, and a capacitance of a parallel resonance capacitor.
The switching power circuit also comprises power factor improving means to improve a power factor by feeding back the switching output, which is obtained in the primary winding, to a rectified current path.
The switching power circuit further comprises a secondary side resonance circuit formed on the secondary side and consisting of a leakage inductance component of the secondary winding of the insulating converter transformer and a capacitance of a secondary side resonance capacitor; DC output voltage generating means formed inclusive of the secondary side resonance circuit and rectifying the input alternating voltage obtained from the secondary winding of the insulating converter transformer, thereby generating a secondary side DC output voltage; and constant voltage control means for executing constant voltage control of the secondary side DC output voltage in accordance with the level of the secondary side DC output voltage.
The power factor improving means consists at least of a tertiary winding coiled for the insulating converter transformer in such a manner as to wind up the primary winding; a resonance capacitor inserted for feeding back, to the rectified current path via the capacitance thereof, the switching output transferred from the primary winding to the tertiary winding; a switching element inserted in the rectified current path to execute a switching operation in accordance with the switching output fed back to the current path; and an inductor inserted in the rectified current path.
According to the configuration described above, the power factor improving circuit, which is included in the power circuit termed a composite resonance type converter, transfers to the tertiary winding the switching output obtained in the primary winding of the insulating converter transformer, and then feeds back the switching output from the tertiary winding to the rectified current path via the resonance capacitor. In this configuration, the switching output transferred via the tertiary winding serves to produce a period during which the resonance circuit composed of the resonance capacitor and the inductor in the power factor improving means performs its resonance operation.
The above and other features and advantages of the present invention will become apparent from the following description which will be given with reference to the illustrative accompanying drawings.